DS25BR101TSDE/NOPB - NATIONAL SEMICONDUCTOR

LM3405

LM3405 1.6MHz, 1A Constant Current Buck Regulator for Powering LEDs

Literature Number: SNVS429A

February 2007

LM3405

1.6MHz, 1A Constant Current Buck Regulator for Powering

LEDs

General Description

Integrated with a 1A power switch, the LM3405 is a current-

mode control switching buck regulator designed to provide a

simple, high efficiency solution for driving high power LEDs.

With a 0.205V reference voltage feedback control to minimize

power dissipation, an external resistor sets the current as

needed for driving various types of LEDs. Switching frequen-

cy is internally set to 1.6MHz, allowing small surface mount

inductors and capacitors to be used. The LM3405 utilizes

current-mode control and internal compensation offering

ease of use and predictable, high performance regulation

over a wide range of operating conditions. Additional features

include user accessible EN/DIM pin for enabling and PWM

dimming of LEDs, thermal shutdown, cycle-by-cycle current

limit and over-current protection.

Features

■VIN operating range of 3V to 15V

■Thin SOT23-6 package

■1.6MHz switching frequency

■300mΩ NMOS switch

■40nA shutdown current at VIN = 5V

■EN/DIM input for enabling and PWM dimming of LEDs

■Internally compensated current-mode control

■Cycle-by-cycle current limit

■Input voltage UVLO

■Over-current protection

■Thermal shutdown

Applications

■LED Driver

■Constant Current Source

■Industrial Lighting

■LED Flashlights

Typical Application Circuit

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Efficiency vs LED Current (VIN = 5V)

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LM3405 1.6MHz, 1A Constant Current Buck Regulator for Powering LEDs

Connection Diagrams

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6-Lead TSOT

NS Package Number MK06A

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Pin 1 Identification

Ordering Information

Part Number Package Type NS Package Drawing Package Marking Supplied As

LM3405XMK TSOT-6 MK06A SPNB 1000 Units on Tape and Reel

LM3405XMKX SPNB 3000 Units on Tape and Reel

*NOPB versions are available

Pin Descriptions

Pin(s) Name Application Information

1 BOOST Voltage at this pin drives the internal NMOS power switch. A bootstrap capacitor is

connected between the BOOST and SW pins.

2 GND Signal and Power ground pin. Place the LED current-setting resistor as close as possible

to this pin for accurate current regulation.

3 FB Feedback pin. Connect an external resistor from FB to GND to set the LED Current.

4 EN/DIM

Enable control input. Logic high enables operation. Toggling this pin with a periodic logic

square wave of varying duty cycle at different frequencies controls the brightness of LEDs.

Do not allow this pin to float or be greater than VIN + 0.3V.

5 VIN Input supply voltage. Connect a bypass capacitor locally from this pin to GND.

6 SW Switch pin. Connect this pin to the inductor, catch diode, and bootstrap capacitor.

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LM3405

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VIN -0.5V to 20V

SW Voltage -0.5V to 20V

Boost Voltage -0.5V to 26V

Boost to SW Voltage -0.5V to 6.0V

FB Voltage -0.5V to 3.0V

EN/DIM Voltage -0.5V to (VIN + 0.3V)

Junction Temperature 150°C

ESD Susceptibility (Note 2) 2kV

Storage Temperature -65°C to +150°C

Soldering Information

Infrared/Convection Reflow (15sec) 220°C

Operating Ratings (Note 1)

VIN 3V to 15V

EN/DIM voltage -0.5V to (VIN + 0.3V)

Boost to SW Voltage 2.5V to 5.5V

Junction Temperature Range -40°C to +125°C

Thermal Resistance θJA (Note 3) 118°C/W

Electrical Characteristics Unless otherwise specified, VIN = 12V. Limits in standard type are for TJ = 25°C only;

limits in boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are

guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm, and are

provided for reference purposes only.

Symbol Parameter Conditions Min Typ Max Units

VFB Feedback Voltage 0.188 0.205 0.220 V

ΔVFB/(ΔVINxVFB)Feedback Voltage Line Regulation VIN = 3V to 15V 0.01 %/V

IFB Feedback Input Bias Current Sink/Source 10 250 nA

UVLO

Under-voltage Lockout VIN Rising 2.74 2.95 V

Under-voltage Lockout VIN Falling 1.9 2.3 V

UVLO Hysteresis 0.44 V

fSW Switching Frequency 1.2 1.6 1.9 MHz

DMAX Maximum Duty Cycle VFB = 0V 85 94 %

RDS(ON) Switch ON Resistance VBOOST - VSW = 3V 300 600 mΩ

ICL Switch Current Limit VBOOST - VSW = 3V, VIN = 3V 1.2 2.0 2.8 A

Quiescent Current Switching, VFB = 0.195V 1.8 2.8 mA

Quiescent Current (Shutdown) VEN/DIM = 0V 0.3 µA

VEN/DIM_TH

Enable Threshold Voltage VEN/DIM Rising 1.8 V

Shutdown Threshold Voltage VEN/DIM Falling 0.4 V

IEN/DIM EN/DIM Pin Current Sink/Source 0.01 µA

ISW Switch Leakage VIN = 15V 0.1 µA

Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings define the conditions under which the device

is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.

Note 2: Human body model, 1.5kΩ in series with 100pF.

Note 3: Thermal shutdown will occur if the junction temperature (TJ) exceeds 165°C. The maximum allowable power dissipation (PD) at any ambient temperature

(TA) is PD = (TJ(MAX) – TA)/θJA . This number applies to packages soldered directly onto a 3" x 3" PC board with 2oz. copper on 4 layers in still air. For a 2 layer

board using 1 oz. copper in still air, θJA = 204°C/W.

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LM3405

Typical Performance Characteristics Unless otherwise specified, VIN = 12V, VBOOST - VSW = 5V and

TA = 25°C.

Efficiency vs LED Current

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Efficiency vs Input Voltage (IF = 1A)

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Efficiency vs Input Voltage (IF = 0.7A)

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Efficiency vs Input Voltage (IF = 0.35A)

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VFB vs Temperature

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Oscillator Frequency vs Temperature

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LM3405

Current Limit vs Temperature

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RDS(ON) vs Temperature (VBOOST - VSW = 3V)

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Quiescent Current vs Temperature

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Startup Response to EN/DIM Signal

(VIN = 15V, IF = 0.2A)

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LM3405

Block Diagram

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FIGURE 1. Simplified Block Diagram

Application Information

THEORY OF OPERATION

The LM3405 is a PWM, current-mode control switching buck

regulator designed to provide a simple, high efficiency solu-

tion for driving LEDs with a preset switching frequency of

1.6MHz. This high frequency allows the LM3405 to operate

with small surface mount capacitors and inductors, resulting

in LED drivers that need only a minimum amount of board

space. The LM3405 is internally compensated, simple to use,

and requires few external components.

The following description of operation of the LM3405 will refer

to the Simplified Block Diagram (Figure 1) and to the wave-

forms in Figure 2. The LM3405 supplies a regulated output

current by switching the internal NMOS power switch at con-

stant frequency and variable duty cycle. A switching cycle

begins at the falling edge of the reset pulse generated by the

internal oscillator. When this pulse goes low, the output con-

trol logic turns on the internal NMOS power switch. During this

on-time, the SW pin voltage (VSW) swings up to approximately

VIN, and the inductor current (IL) increases with a linear slope.

IL is measured by the current sense amplifier, which gener-

ates an output proportional to the switch current. The sense

signal is summed with the regulator’s corrective ramp and

compared to the error amplifier’s output, which is proportional

to the difference between the feedback voltage and VREF.

When the PWM comparator output goes high, the internal

power switch turns off until the next switching cycle begins.

During the switch off-time, inductor current discharges

through the catch diode D1, which forces the SW pin to swing

below ground by the forward voltage (VD1) of the catch diode.

The regulator loop adjusts the duty cycle (D) to maintain a

constant output current (IF) through the LED, by forcing FB

pin voltage to be equal to VREF (0.205V).

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FIGURE 2. SW Pin Voltage and Inductor Current

Waveforms of LM3405

BOOST FUNCTION

Capacitor C3 and diode D2 in Figure 1 are used to generate

a voltage VBOOST. The voltage across C3, VBOOST - VSW, is

the gate drive voltage to the internal NMOS power switch. To

properly drive the internal NMOS switch during its on-time,

VBOOST needs to be at least 2.5V greater than VSW. Large

value of VBOOST - VSW is recommended to achieve better ef-

ficiency by minimizing both the internal switch ON resistance

(RDS(ON)), and the switch rise and fall times. However,

VBOOST - VSW should not exceed the maximum operating limit

of 5.5V.

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LM3405

When the LM3405 starts up, internal circuitry from VIN sup-

plies a 20mA current to the BOOST pin, flowing out of the

BOOST pin into C3. This current charges C3 to a voltage suf-

ficient to turn the switch on. The BOOST pin will continue to

source current to C3 until the voltage at the feedback pin is

greater than 123mV.

There are various methods to derive VBOOST:

1. From the input voltage (VIN)

2. From the output voltage (VOUT)

3. From a shunt or series zener diode

4. From an external distributed voltage rail (VEXT)

The first method is shown in the Simplified Block Diagram of

Figure 1. Capacitor C3 is charged via diode D2 by VIN. During

a normal switching cycle, when the internal NMOS power

switch is off (TOFF) (refer to Figure 2), VBOOST equals VIN mi-

nus the forward voltage of D2 (VD2), during which the current

in the inductor (L1) forward biases the catch diode D1 (VD1).

Therefore the gate drive voltage stored across C3 is:

VBOOST - VSW = VIN - VD2 + VD1

When the NMOS switch turns on (TON), the switch pin rises

to:

VSW = VIN – (RDS(ON) x IL)

Since the voltage across C3 remains unchanged, VBOOST is

forced to rise thus reverse biasing D2. The voltage at

VBOOST is then:

VBOOST = 2VIN – (RDS(ON) x IL) – VD2 + VD1

Depending on the quality of the diodes D1 and D2, the gate

drive voltage in this method can be slightly less or larger than

the input voltage VIN. For best performance, ensure that the

variation of the input supply does not cause the gate drive

voltage to fall outside the recommended range:

2.5V < VIN - VD2 + VD1 < 5.5V

The second method for deriving the boost voltage is to con-

nect D2 to the output as shown in Figure 3. The gate drive

voltage in this configuration is:

VBOOST - VSW = VOUT – VD2 + VD1

Since the gate drive voltage needs to be in the range of 2.5V

to 5.5V, the output voltage VOUT should be limited to a certain

range. For the calculation of VOUT, see OUTPUT VOLTAGE

section.

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FIGURE 3. VBOOST derived from VOUT

The third method can be used in the applications where both

VIN and VOUT are greater than 5.5V. In these cases, C3 cannot

be charged directly from these voltages; instead C3 can be

charged from VIN or VOUT minus a zener voltage (VD3) by

placing a zener diode D3 in series with D2 as shown in Figure

4. When using a series zener diode from the input, the gate

drive voltage is VIN - VD3 - VD2 + VD1.

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FIGURE 4. VBOOST derived from VIN through a Series

Zener

An alternate method is to place the zener diode D3 in a shunt

configuration as shown in Figure 5. A small 350mW to

500mW, 5.1V zener in a SOT-23 or SOD package can be

used for this purpose. A small ceramic capacitor such as a

6.3V, 0.1µF capacitor (C5) should be placed in parallel with

the zener diode. When the internal NMOS switch turns on, a

pulse of current is drawn to charge the internal NMOS gate

capacitance. The 0.1µF parallel shunt capacitor ensures that

the VBOOST voltage is maintained during this time. Resistor R2

should be chosen to provide enough RMS current to the zener

diode and to the BOOST pin. A recommended choice for the

zener current (IZENER) is 1mA. The current IBOOST into the

BOOST pin supplies the gate current of the NMOS power

switch. It reaches a maximum of around 3.6mA at the highest

gate drive voltage of 5.5V over the LM3405 operating range.

For the worst case IBOOST, increase the current by 50%. In

that case, the maximum boost current will be:

IBOOST-MAX = 1.5 x 3.6mA = 5.4mA

R2 will then be given by:

R2 = (VIN - VZENER) / (IBOOST_MAX + IZENER)

For example, let VIN = 12V, VZENER = 5V, IZENER = 1mA, then:

R2 = (12V - 5V) / (5.4mA + 1mA) = 1.09kΩ

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FIGURE 5. VBOOST derived from VIN through a Shunt Zener

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LM3405

The fourth method can be used in an application which has

an external low voltage rail, VEXT. C3 can be charged through

D2 from VEXT, independent of VIN and VOUT voltage levels.

Again for best performance, ensure that the gate drive volt-

age, VEXT - VD2 + VD1, falls in the range of 2.5V to 5.5V.

SETTING THE LED CURRENT

LM3405 is a constant current buck regulator. The LEDs are

connected between VOUT and FB pin as shown in the Typical

Application Circuit. The FB pin is at 0.205V in regulation and

therefore the LED current IF is set by VFB and the resistor R1

from FB to ground by the following equation:

IF = VFB / R1

IF should not exceed the 1A current capability of LM3405 and

therefore R1 minimum must be approximately 0.2Ω. IF should

also be kept above 200mA for stable operation, and therefore

R1 maximum must be approximately 1Ω. If average LED cur-

rents less than 200mA are desired, the EN/DIM pin can be

used for PWM dimming. See LED PWM DIMMING section.

OUTPUT VOLTAGE

The output voltage is primarily determined by the number of

LEDs (n) connected from VOUT to FB pin and therefore VOUT

can be written as :

VOUT = ((n x VF) + VFB)

where VF is the forward voltage of one LED at the set LED

current level (see LED manufacturer datasheet for forward

characteristics curve).

ENABLE MODE / SHUTDOWN MODE

The LM3405 has both enable and shutdown modes that are

controlled by the EN/DIM pin. Connecting a voltage source

greater than 1.8V to the EN/DIM pin enables the operation of

LM3405, while reducing this voltage below 0.4V places the

part in a low quiescent current (0.3µA typical) shutdown

mode. There is no internal pull-up on EN/DIM pin, therefore

an external signal is required to initiate switching. Do not allow

this pin to float or rise to 0.3V above VIN. It should be noted

that when the EN/DIM pin voltage rises above 1.8V while the

input voltage is greater than UVLO, there is a finite delay be-

fore switching starts. During this delay the LM3405 will go

through a power on reset state after which the internal soft-

start process commences. The soft-start process limits the

inrush current and brings up the LED current (IF) in a smooth

and controlled fashion. The total combined duration of the

power on reset delay, soft-start delay and the delay to fully

establish the LED current is in the order of 100µs (refer to

Figure 11).

The simplest way to enable the operation of LM3405 is to

connect the EN/DIM pin to VIN which allows self start-up of

LM3405 whenever the input voltage is applied. However,

when an input voltage of slow rise time is used to power the

application and if both the input voltage and the output voltage

are not fully established before the soft-start time elapses, the

control circuit will command maximum duty cycle operation of

the internal power switch to bring up the output voltage rapid-

ly. When the feedback pin voltage exceeds 0.205V, the duty

cycle will have to reduce from the maximum value according-

ly, to maintain regulation. It takes a finite amount of time for

this reduction of duty cycle and this will result in a spike in LED

current for a short duration as shown in Figure 6. In applica-

tions where this LED current overshoot is undesirable, EN/

DIM pin voltage can be delayed with respect to VIN such that

VIN is fully established before the EN/DIM pin voltage reaches

the enable threshold. This delay can be implemented by a

simple Ra-Ca network as shown in Figure 7. The effect of

adding this Ra-Ca network on the LED current is shown in

Figure 8. For a fast rising input voltage (200µs for example),

there is no need to delay the EN/DIM signal since soft-start

can smoothly bring up the LED current as shown in Figure

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FIGURE 6. Startup Response to VIN with 5ms rise time

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FIGURE 7. EN/DIM delayed with respect to VIN

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FIGURE 8. Startup Response to VIN with EN/DIM delayed

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LM3405

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FIGURE 9. Startup Response to VIN with 200µs rise time

LED PWM DIMMING

The LED brightness can be controlled by applying a periodic

pulse signal to the EN/DIM pin and varying its frequency and/

or duty cycle. This so-called PWM dimming method controls

the average light output by pulsing the LED current between

the set value and zero. A logic high level at the EN/DIM pin

turns on the LED current whereas a logic low level turns off

the LED current. Figure 10 shows a typical LED current wave-

form in PWM dimming mode. As explained in the previous

section, there is approximately a 100µs delay from the EN/

DIM signal going high to fully establishing the LED current as

shown in Figure 11. This 100µs delay sets a maximum fre-

quency limit for the driving signal that can be applied to the

EN/DIM pin for PWM dimming. Figure 12 shows the average

LED current versus duty cycle of PWM dimming signal for

various frequencies. The applicable frequency range to drive

LM3405 for PWM dimming is from 100Hz to 5kHz. The dim-

ming ratio reduces drastically when the applied PWM dim-

ming frequency is greater than 5kHz.

20178966

FIGURE 10. PWM Dimming of LEDs using the EN/DIM Pin

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FIGURE 11. Startup Response to EN/DIM with IF = 1A

20178983

FIGURE 12. Average LED Current versus Duty Cycle of

PWM Dimming Signal at EN/DIM Pin

UNDER-VOLTAGE LOCKOUT

Under-voltage lockout (UVLO) prevents the LM3405 from op-

erating until the input voltage exceeds 2.74V (typical). The

UVLO threshold has approximately 440mV of hysteresis, so

the part will operate until VIN drops below 2.3V (typical). Hys-

teresis prevents the part from turning off during power up if

VIN is non-monotonic.

CURRENT LIMIT

The LM3405 uses cycle-by-cycle current limit to protect the

internal power switch. During each switching cycle, a current

limit comparator detects if the power switch current exceeds

2.0A (typical), and turns off the switch until the next switching

cycle begins.

OVER-CURRENT PROTECTION

The LM3405 has a built in over-current comparator that com-

pares the FB pin voltage to a threshold voltage that is 60%

higher than the internal reference VREF. Once the FB pin volt-

age exceeds this threshold level (typically 328mV), the inter-

nal NMOS power switch is turned off, which allows the

feedback voltage to decrease towards regulation. This

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LM3405

threshold provides an upper limit for the LED current. LED

current overshoot is limited to 328mV/R1 by this comparator

during transients.

THERMAL SHUTDOWN

Thermal shutdown limits total power dissipation by turning off

the internal power switch when the IC junction temperature

exceeds 165°C. After thermal shutdown occurs, the power

switch does not turn on until the junction temperature drops

below approximately 150°C.

Design Guide

INDUCTOR (L1)

The Duty Cycle (D) can be approximated quickly using the

ratio of output voltage (VOUT) to input voltage (VIN):

The catch diode (D1) forward voltage drop and the voltage

drop across the internal NMOS must be included to calculate

a more accurate duty cycle. Calculate D by using the following

formula:

VSW can be approximated by:

VSW = IF x RDS(ON)

The diode forward drop (VD1) can range from 0.3V to 0.7V

depending on the quality of the diode. The lower VD1 is, the

higher the operating efficiency of the converter.

The inductor value determines the output ripple current (ΔiL,

as defined in Figure 2). Lower inductor values decrease the

size of the inductor, but increases the output ripple current.

An increase in the inductor value will decrease the output rip-

ple current. The ratio of ripple current to LED current is

optimized when it is set between 0.3 and 0.4 at 1A LED cur-

rent. This ratio r is defined as:

One must also ensure that the minimum current limit (1.2A)

is not exceeded, so the peak current in the inductor must be

calculated. The peak current (ILPK) in the inductor is calculated

as:

ILPK = IF + ΔiL/2

When the designed maximum output current is reduced, the

ratio r can be increased. At a current of 0.2A, r can be made

as high as 0.7. The ripple ratio can be increased at lighter

loads because the net ripple is actually quite low, and if r re-

mains constant the inductor value can be made quite large.

An equation empirically developed for the maximum ripple

ratio at any current below 2A is:

r = 0.387 x IOUT-0.3667

Note that this is just a guideline.

The LM3405 operates at a high frequency allowing the use of

ceramic output capacitors without compromising transient re-

sponse. Ceramic capacitors allow higher inductor ripple with-

out significantly increasing LED current ripple. See the output

capacitor and feed-forward capacitor sections for more de-

tails on LED current ripple.

Now that the ripple current or ripple ratio is determined, the

inductance is calculated by:

where fSW is the switching frequency and IF is the LED current.

When selecting an inductor, make sure that it is capable of

supporting the peak output current without saturating. Induc-

tor saturation will result in a sudden reduction in inductance

and prevent the regulator from operating correctly. Because

of the operating frequency of LM3405, ferrite based inductors

are preferred to minimize core losses. This presents little re-

striction since the variety of ferrite based inductors is huge.

Lastly, inductors with lower series resistance (DCR) will pro-

vide better operating efficiency. For recommended inductor

selection, refer to Circuit Examples and Recommended In-

ductance Range in Table 1. Note that it is a good practice to

use small inductance value at light load (for example, IF =

0.2A) to increase inductor current ramp signal, such that noise

immunity is improved.

TABLE 1. Recommended Inductance Range

IFInductance Range and Inductor Current Ripple

1.0A

4.7µH-10µH

Inductance 4.7µH 6.8µH 10µH

ΔiL / IF*51% 35% 24%

0.6A

6.8µH-15µH

Inductance 6.8µH 10µH 15µH

ΔiL / IF*58% 40% 26%

0.2A

4.7µH**-22µH

Inductance 10µH 15µH 22µH

ΔiL / IF*119% 79% 54%

*Maximum over full range of VIN and VOUT.

**Small inductance improves stability without causing a significant increase

in LED current ripple.

INPUT CAPACITOR (C1)

An input capacitor is necessary to ensure that VIN does not

drop excessively during switching transients. The primary

specifications of the input capacitor are capacitance, voltage

rating, RMS current rating, and ESL (Equivalent Series In-

ductance). The input voltage rating is specifically stated by

the capacitor manufacturer. Make sure to check any recom-

mended deratings and also verify if there is any significant

change in capacitance at the operating input voltage and the

operating temperature. The input capacitor maximum RMS

input current rating (IRMS-IN) must be greater than:

It can be shown from the above equation that maximum RMS

capacitor current occurs when D = 0.5. Always calculate the

RMS at the point where the duty cycle D, is closest to 0.5. The

ESL of an input capacitor is usually determined by the effec-

tive cross sectional area of the current path. A large leaded

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LM3405

capacitor will have high ESL and a 0805 ceramic chip capac-

itor will have very low ESL. At the operating frequency of the

LM3405, certain capacitors may have an ESL so large that

the resulting inductive impedance (2πfL) will be higher than

that required to provide stable operation. It is strongly recom-

mended to use ceramic capacitors due to their low ESR and

low ESL. A 10µF multilayer ceramic capacitor (MLCC) is a

good choice for most applications. In cases where large ca-

pacitance is required, use surface mount capacitors such as

Tantalum capacitors and place at least a 1µF ceramic capac-

itor close to the VIN pin. For MLCCs it is recommended to use

X7R or X5R dielectrics. Consult capacitor manufacturer

datasheet to see how rated capacitance varies over operating

conditions.

OUTPUT CAPACITOR (C2)

The output capacitor is selected based upon the desired re-

duction in LED current ripple. A 1µF ceramic capacitor results

in very low LED current ripple for most applications. Due to

the high switching frequency, the 1µF capacitor alone (without

feed-forward capacitor C4) can filter more than 90% of the

inductor current ripple for most applications where the sum of

LED dynamic resistance and R1 is larger than 1Ω. Since the

internal compensation is tailored for small output capacitance

with very low ESR, it is strongly recommended to use a ce-

ramic capacitor with capacitance less than 3.3µF.

Given the availability and quality of MLCCs and the expected

output voltage of designs using the LM3405, there is really no

need to review other capacitor technologies. A benefit of ce-

ramic capacitors is their ability to bypass high frequency

noise. A certain amount of switching edge noise will couple

through the parasitic capacitances in the inductor to the out-

put. A ceramic capacitor will bypass this noise. In cases where

large capacitance is required, use Electrolytic or Tantalum

capacitors with large ESR, and verify the loop performance

on bench. Like the input capacitor, recommended multilayer

ceramic capacitors are X7R or X5R. Again, verify actual ca-

pacitance at the desired operating voltage and temperature.

Check the RMS current rating of the capacitor. The maximum

RMS current rating of the capacitor is:

One may select a 1206 size ceramic capacitor for C2, since

its current rating is typically higher than 1A, more than enough

for the requirement.

FEED-FORWARD CAPACITOR (C4)

The feed-forward capacitor (designated as C4) connected in

parallel with the LED string is required to provide multiple

benefits to the LED driver design. It greatly improves the large

signal transient response and suppresses LED current over-

shoot that may otherwise occur during PWM dimming; it also

helps to shape the rise and fall times of the LED current pulse

during PWM dimming thus reducing EMI emission; it reduces

LED current ripple by bypassing some of inductor ripple from

flowing through the LED. For most applications, a 1µF ce-

ramic capacitor is sufficient. In fact, the combination of a 1µF

feed-forward ceramic capacitor and a 1µF output ceramic ca-

pacitor leads to less than 1% current ripple flowing through

the LED. Lower and higher C4 values can be used, but bench

validation is required to ensure the performance meets the

application requirement.

Figure 13 shows a typical LED current waveform during PWM

dimming without feed-forward capacitor. At the beginning of

each PWM cycle, overshoot can be seen in the LED current.

Adding a 1µF feed-forward capacitor can totally remove the

overshoot as shown in Figure 14.

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FIGURE 13. PWM Dimming without Feed-Forward

Capacitor

20178970

FIGURE 14. PWM Dimming with a 1µF Feed-Forward

Capacitor

CATCH DIODE (D1)

The catch diode (D1) conducts during the switch off-time. A

Schottky diode is required for its fast switching time and low

forward voltage drop. The catch diode should be chosen such

that its current rating is greater than:

ID1 = IF x (1-D)

The reverse breakdown rating of the diode must be at least

the maximum input voltage plus appropriate margin. To im-

prove efficiency, choose a Schottky diode with a low forward

voltage drop.

BOOST DIODE (D2)

A standard diode such as the 1N4148 type is recommended.

For VBOOST circuits derived from voltages less than 3.3V, a

small-signal Schottky diode is recommended for better effi-

ciency. A good choice is the BAT54 small signal diode.

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LM3405

BOOST CAPACITOR (C3)

A 0.01µF ceramic capacitor with a voltage rating of at least

6.3V is sufficient. The X7R and X5R MLCCs provide the best

performance.

POWER LOSS ESTIMATION

The main power loss in LM3405 includes three basic types of

loss in the internal power switch: conduction loss, switching

loss, and gate charge loss. In addition, there is loss associ-

ated with the power required for the internal circuitry of IC.

The conduction loss is calculated as:

If the inductor ripple current is fairly small (for example, less

than 40%) , the conduction loss can be simplified to:

PCOND = IF2 x RDS(ON) x D

The switching loss occurs during the switch on and off tran-

sition periods, where voltage and current overlap resulting in

power loss. The simplest means to determine this loss is to

empirically measure the rise and fall times (10% to 90%) of

the voltage at the switch pin.

Switching power loss is calculated as follows:

PSW = 0.5 x VIN x IF x fSW x ( TRISE + TFALL )

The gate charge loss is associated with the gate charge QG

required to drive the switch:

PG = fSW x VIN x QG

The power loss required for operation of the internal circuitry:

PQ = IQ x VIN

IQ is the quiescent operating current, and is typically around

1.8mA for the LM3405.

The total power loss in the IC is:

PINTERNAL = PCOND + PSW + PG + PQ

An example of power losses for a typical application is shown

in Table 2:

TABLE 2. Power Loss Tabulation

Conditions Power loss

VIN 12V

VOUT 4.1V

IOUT 1.0A

VD1 0.45V

RDS(ON) 300mΩPCOND 111mW

fSW 1.6MHz

TRISE 18ns PSW 288mW

TFALL 12ns

IQ1.8mA PQ22mW

QG1.4nC PG27mW

D is calculated to be 0.37

Σ ( PCOND + PSW + PQ + PG ) = PINTERNAL

PINTERNAL = 448mW

PCB Layout Considerations

When planning layout there are a few things to consider when

trying to achieve a clean, regulated output. The most impor-

tant consideration when completing the layout is the close

coupling of the GND connections of the input capacitor C1

and the catch diode D1. These ground ends should be close

to one another and be connected to the GND plane with at

least two through-holes. Place these components as close to

the IC as possible. The next consideration is the location of

the GND connection of the output capacitor C2, which should

be near the GND connections of C1 and D1.

There should be a continuous ground plane on the bottom

layer of a two-layer board except under the switching node

island.

The FB pin is a high impedance node and care should be

taken to make the FB trace short to avoid noise pickup that

causes inaccurate regulation. The LED current setting resis-

tor R1 should be placed as close as possible to the IC, with

the GND of R1 placed as close as possible to the GND of the

IC. The VOUT trace to LED anode should be routed away from

the inductor and any other traces that are switching.

High AC currents flow through the VIN, SW and VOUT traces,

so they should be as short and wide as possible. Radiated

noise can be decreased by choosing a shielded inductor.

The remaining components should also be placed as close

as possible to the IC. Please see Application Note AN-1229

for further considerations and the LM3405 demo board as an

example of a four-layer layout.

www.national.com 12

LM3405

LM3405 Circuit Examples

20178942

FIGURE 15. VBOOST derived from VIN

( VIN = 5V, IF = 1A )

Bill of Materials for Figure 15

Part ID Part Value Part Number Manufacturer

U1 1A LED Driver LM3405 National Semiconductor

C1, Input Cap 10µF, 6.3V, X5R C3216X5R0J106M TDK

C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata

C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX

C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata

D1, Catch Diode Schottky, 0.37V at 1A, VR = 10V MBRM110LT1G ON Semiconductor

D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor

L1 4.7µH, 1.6A SLF6028T-4R7M1R6 TDK

R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay

LED1 1A, White LED LXHL-PW09 Lumileds

13 www.national.com

LM3405

20178943

FIGURE 16. VBOOST derived from VOUT

( VIN = 12V, IF = 1A )

Bill of Materials for Figure 16

Part ID Part Value Part Number Manufacturer

U1 1A LED Driver LM3405 National Semiconductor

C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic

C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata

C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX

C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata

D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay

D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor

L1 4.7µH, 1.6A SLF6028T-4R7M1R6 TDK

R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay

LED1 1A, White LED LXHL-PW09 Lumileds

www.national.com 14

LM3405

20178944

FIGURE 17. VBOOST derived from VIN through a Shunt Zener Diode (D3)

( VIN = 15V, IF = 1A )

Bill of Materials for Figure 17

Part ID Part Value Part Number Manufacturer

U1 1A LED Driver LM3405 National Semiconductor

C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic

C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata

C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX

C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata

C5, Shunt Cap 0.1µF, 16V, X7R GRM219R71C104KA01D Murata

D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay

D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor

D3, Zener Diode 4.7V, 350mW, SOT-23 BZX84C4V7 Fairchild

L1 6.8µH, 1.5A SLF6028T-6R8M1R5 TDK

R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay

R2 1.91kΩ, 1% CRCW08051K91FKEA Vishay

LED1 1A, White LED LXHL-PW09 Lumileds

15 www.national.com

LM3405

20178949

FIGURE 18. VBOOST derived from VIN through a Series Zener Diode (D3)

( VIN = 15V, IF = 1A )

Bill of Materials for Figure 18

Part ID Part Value Part Number Manufacturer

U1 1A LED Driver LM3405 National Semiconductor

C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic

C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata

C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX

C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata

D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay

D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor

D3, Zener Diode 11V, 350mW, SOT-23 BZX84C11 Fairchild

L1 6.8µH, 1.5A SLF6028T-6R8M1R5 TDK

R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay

LED1 1A, White LED LXHL-PW09 Lumileds

www.national.com 16

LM3405

20178950

FIGURE 19. VBOOST derived from VOUT through a Series Zener Diode (D3)

( VIN = 15V, IF = 1A )

Bill of Materials for Figure 19

Part ID Part Value Part Number Manufacturer

U1 1A LED Driver LM3405 National Semiconductor

C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic

C2, Output Cap 1µF, 16V, X7R GRM319R71A105KC01D Murata

C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX

C4, Feedforward Cap 1µF, 16V, X7R GRM319R71A105KC01D Murata

D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay

D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor

D3, Zener Diode 3.9V, 350mW, SOT-23 BZX84C3V9 Fairchild

L1 6.8µH, 1.5A SLF6028T-6R8M1R5 TDK

R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay

LED1 1A, White LED LXHL-PW09 Lumileds

LED2 1A, White LED LXHL-PW09 Lumileds

17 www.national.com

LM3405

Physical Dimensions inches (millimeters) unless otherwise noted

6-Lead TSOT Package

NS Package Number MK06A

www.national.com 18

LM3405

Notes

19 www.national.com

LM3405

Notes

LM3405 1.6MHz, 1A Constant Current Buck Regulator for Powering LEDs

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